System and method for adaptive broadcast radar system

ABSTRACT

An adaptive broadcast radar system for tracking targets is disclosed, the radar system includes a transmitter having sub-apertures and a receiver having sub-apertures. The transmitter sub-apertures generate and code a signal waveform. The signal waveform is coded with data about the transmitter, including the degrees of freedom. The receiver receives signals comprising direct path signals and scattered signals correlating to the signal waveforms from the transmitter. The receiver includes a signal processor that regenerates a transmit beam for the coded data, delay, and doppler information from the received signals. The signal processor generates data quads encapsulating the information.

CROSS REFERENCE TO RELATED APPLICATIONS

[0001] This application claims benefit of U.S. Provisional PatentApplication No. 60/253,095 entitled “Adaptive Broadcast Radar,” filedNov. 28, 2000, which is hereby incorporated by reference.

BACKGROUND OF THE INVENTION

[0002] 1. Field of the Invention

[0003] The present invention relates to a radar system and method, andmore particularly, to a system and method for performing adaptivebroadcast radar operations.

[0004] 2. Discussion of the Related Art

[0005] Radar systems may be represented by a bistatic or multistaticradar system. A multistatic radar system has many receivers that areseparated from one or more transmitters. The radiated signal from atransmitter arrives at a receiver via two separate paths. One path maybe a direct path from the transmitter to the receiver, and the otherpath may be a target path that includes an indirect path from thetransmitter to a target to the receiver. Measurements may include atotal path length, or transit time, of the target path signal, the angleof arrival of the target path signal, and the frequency of the directand target path signals. A difference in frequency may be detected ifthe target is in motion according to a doppler effect.

[0006] Knowledge of the transmitted signal is desirable at the receiverif information is to be extracted from the target path signal. Thetransmitted frequency is desired to determine the doppler frequencyshift. A time or phase reference also is desired if the total scatteredpath length is to be determined. The frequency reference may be obtainedfrom the direct signal. The time reference also may be obtained from thedirect signal provided the distance between the transmitter and thereceiver is known.

[0007] Multistatic radar systems may be capable of determining thepresence of a target within the coverage of the radar, the location ofthe target position, and a velocity component, or doppler, relative tothe radar. The process of locating the target position may include ameasurement of a distance and the angle of arrival. The measurement ofdistance relative to the receiving site may desire both the angle ofarrival at the receiving site and the distance between transmitter andreceiver. If the direct signal is available, it may be used as areference signal to extract the doppler frequency shift.

[0008] Known radar systems may transmit a signal beam in a specificdirection to search for targets. Once a target has been detected, thebeam may be directed to follow the target. The receiver may receivescattered signals reflected off the target. By knowing the transmitterbeam parameters, the receiver may perform operations to determine thetarget parameters, as disclosed above.

[0009] Future airborne radar systems may operate in a difficultenvironment where the detection of small and maneuverable targets mayoccur against a strong clutter background and jamming operations.Directed beams of energy from transmitters may be susceptible to jammingcountermeasures and detection. Power aperture increases may not beeffective to overcome these limitations and countermeasures againstradar detection. Thus, future systems may desire increase sensitivitywithout increasing power requirements. This condition may be applicableespecially to radar systems where the transmitter power is notcontrolled by the receiving party.

[0010] Mobile radar systems often operate in the presence of jamminginterference and monostatic clutter that produced naturally by groundreflections. Difficulties may arise if both the transmitter and receiverare in motion, such as an airborne radar systems. When both thetransmitter and receiver of a radar system are in motion, the rank ofthe clutter covariance may be increased. An increased number of degreeof freedom in the receiver system may be needed to achieve a specifiedlevel of clutter suppression. Thus, a transmitter or receiver in motionmay increase the clutter interference with a signal, or increase thecomplexity within the receiver in accounting for the increased degreesof freedom.

SUMMARY OF THE INVENTION

[0011] Accordingly, the present invention is directed to multistaticradar applications and signal processing. Thus, a system and method foradaptive broadcast radar operations is disclosed herein.

[0012] According to a disclosed embodiment, a method for formattingreceived data within an adaptive broadcast radar system having atransmitter comprising sub-apertures and a receiver comprisingsub-apertures is disclosed. The data is received at the receiver. Themethod includes providing an estimate for a delay of scattered signalcomponents within the received data. The method also includes generatingan index for the estimate. The index may include a transmitter elementnumber and a receiver element number. The method also includesgenerating a data quad for the index. The method also includesestimating a measurement covariance and a weight vector for the dataquad. The data quad is reformatted with the measurement covariance andthe weight vector.

[0013] According to another disclosed embodiment, a method for obtainingtarget parameters within an adaptive broadcast radar system isdisclosed. The method includes coding information about a signalwaveform generated by a transmitter having sub-apertures. The methodalso includes receiving a received signal at a receiver havingsub-apertures corresponding to the sub-apertures of the transmitter. Thesignal correlates to the signal waveform. The method also includesdecoding information about the signal waveform from the received signal.The method also includes determining a data quad from the information.The data quad may include degrees of freedom associated with thetransmitter.

[0014] According to another disclosed embodiment, a method forgenerating a sensor signal for a received signal within an adaptivebroadcast radar system. The method includes defining a clutter componentfor the received signal at a receiver. The clutter component comprises adirect path signal and a scattered signal. The method also includesdefining a channel transfer function. The method also includesgenerating a sampled version of the received signal according to thechannel transfer function at a sample time. The method also includesdetermining a batch of data from the sampled version for a sub-apertureof the receiver at the sample time. The method also includes indexingthe batch of data into the sensor signal model.

[0015] According to another embodiment, a method for transmitting asignal waveform from a transmitter within an adaptive broadcast radarsystem is disclosed. The transmitter comprises at least onesub-aperture. The method includes generating the signal waveform at theat least one sub-aperture. The method also includes coding the signalwaveform at the at least one sub-aperture. The signal waveform is codedwith the transmitter data. The method also includes phase shifting thesignal waveform at the at least one sub-aperture. The method alsoincludes transmitting the coded signal waveform from an array elementcoupled to the sub-aperture according to the phase shifting.

[0016] According to another embodiment, a method for performing radaroperation within an adaptive broadcast radar system is disclosed. Theradar system includes a transmitter having a first plurality ofsub-apertures and a receiver having a second plurality of sub-apertures.The method includes encoding data on a signal waveform at a transmitter.The data includes a number for said sub-apertures of the transmitter anddegrees of freedom for the transmitter. The method also includescontinuously transmitting the signal waveform. The method also includesdetermining a delay value and a doppler value for received signals atsaid receiver. The received signals comprise direct and scatteredsignals of the signal waveform. The method also includes regenerating atransmit signal beam correlating to the signal waveform from the data,the delay value, and the doppler value.

[0017] According to another embodiment, an adaptive broadcast radarsystem is disclosed. The radar system includes a transmitter comprisinga first plurality of sub-apertures. Each sub-aperture codes a signalwaveform with data. The radar system also includes a receiver comprisinga second plurality of sub-apertures coupled to a signal processor,wherein the signal processor generates a transmit beam signal accordingto the data within each signal waveform.

[0018] Additional features and advantages of the invention will be setforth in the description which follows, and in part will be apparentfrom the description, or maybe learned by practice of the invention. Theobjectives and other advantages of the invention will be realized andattained by the structure particularly pointed out in the writtendescription and claims hereof as well as the appended drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0019] The accompanying drawings, which is included to provide furtherunderstanding of the invention and is incorporated in and constitutes apart of this specification, illustrates embodiments of the presentinvention and together with the description serves to explain theprinciples of the invention. In the drawings:

[0020]FIG. 1 illustrates a block diagram of an adaptive broadcast radarsystem for detecting and tracking a target in accordance with anembodiment of the present invention;

[0021]FIG. 2 illustrates a transmitter within an adaptive broadcastradar system in accordance with an embodiment of the present invention;

[0022]FIG. 3 illustrates a receiver within an adaptive broadcast radarsystem in accordance with an embodiment of the present invention;

[0023]FIG. 4 illustrates a channel demultiplexer in accordance with anembodiment of the present invention;

[0024]FIG. 5 illustrates a signal processor in accordance with anembodiment of the present invention;

[0025]FIG. 6 illustrates a flowchart for formatting received data inaccordance with an embodiment of the present invention;

[0026]FIG. 7 illustrates a flowchart for generating a sensor signalmodel in accordance with an embodiment of the present invention; and

[0027]FIG. 8 illustrates a flowchart for performing radar operationswithin an adaptive broadcast radar system in accordance with anembodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0028] Reference will now be made in detail to the preferred embodimentsof the present invention, examples of which are illustrated in theaccompanying drawings.

[0029]FIG. 1 depicts a block diagram of an adaptive broadcast radarsystem for detecting and tracking a target in accordance with anembodiment of the present invention. Radar detection system 10 includesa receiving system 100 to track one or more targets of interest 150 byexploiting signals from a plurality of transmitters 110, 112, and 114.Radar detection system 10 also may be known as an adaptive broadcastradar system.

[0030] Receiving system 100 represents a family of multi-static widearea target surveillance sensors. Receiving system 100 system exploitcontinuous wave (“CW”) electromagnetic energy. Preferably, receivingsystem 100 may receive transmissions from a plurality of transmitters110, 112, and 114. Preferred embodiments of the transmitters for use inan adaptive broadcast radar system are disclosed below in greaterdetail. Transmitters 110, 112, and 114, however, may include any device,system or means to transmit uncontrolled signals.

[0031] Transmitters 110, 112, and 114 may transmit widebandelectromagnetic energy transmissions in all directions. Some of thesetransmissions are reflected by one or more targets of interest 150 andreceived by PCL system 100. For example, reflected transmission 130 maybe reflected by target 150 and received by receiving system 100.Further, with regard to transmitter 114, reference transmission 140 maybe received directly by receiving system 100. Receiving system 100 maycompare reference transmission 140 and reflected transmission 130 todetermine positional information about one or more targets of interest150. Reference transmission 140 also may be known as a direct pathsignal. Reflected transmission 130 also may be known as a target pathsignal. Positional information may include any information relating to aposition of target 150, including location, velocity, and accelerationfrom determining a time difference of arrival (“TDOA”), a frequencydifference of arrival (“FDOA”) and an angle of arrival (“AOA”).

[0032] Receiving system 100 may comprise different components, includingreceiver 102 and processing unit 104. According to the disclosedembodiments, transmitter 111 may be a transmitter array, while receiver102 may be a receiver array. Transmitter 111 may include a plurality ofelements such that each element transmits an independent signal. Thesignals may comprise orthogonal or pseudo- orthogonal signals. Each ofthe plurality of elements for the transmitter and the receiver maycomprise a dipole with a back-plane. Transmitter 111 may be in motion,and not at a fixed position.

[0033] Receiver 102 may be a moving receiver array that includes aplurality of elements such that each element is configured to receive ascattered signal. Further, receiver 102 is configured to receive a setof information relating to the independent signals of each of the firstplurality of elements. The radar functions of the disclosed embodimentsmay be performed using information received by any receiver withsuitable receive equipment and knowledge of the transmitter waveformcodes. Receiver 102 may be in motion, and not at a fixed position.Further, there may be a plurality of receivers, with all the receiversin motion. The receivers may not be coupled together, or incommunication, so as to act independently of each other. For example,each receiver may be on moving vehicles within a certain area ofemphasis. The receivers within the area may receive the transmittedsignals continuously from the transmitters, such as transmitter 111.

[0034] Processing unit 104 is configured to receive information fromreceiver 102 and to determine target 150 location based on the scatteredsignals and the set of information relating to the independent signalsof each of the first plurality of elements. Processing unit 104, alongwith receiver 102, may be an adaptive array used in conjunction with aset of antennae coupled to receiving system 100 to provide a versatileform of spatial filtering. Processing unit 104 may combine spatialsamples of a propagating field with signals 140 with a variable set ofweights. The weights may be chosen to reject interfering signals andnoise. Specifically, the spatial filtering capability of the array mayfacilitate cancellation of hostile jamming signals and suppression ofclutter.

[0035] Processing unit 104 may reformulate each of the independentsignals provided by receiver 102. Accordingly, each processing unit 104may form independently all potential beams generated by transmitter 111,or any subset of beams generated by transmitter 111. Because processingunit 104 does not control transmitter 111, a single transmitter may beutilized by multiple receiver/processing unit combinations. Thus,signals from a single transmitter may be recreated at each receiver,independently of the transmitter and the other receivers.

[0036] Processing unit 104 may be located physically with receiver 102such that one processing unit is collocated or integral with eachreceiver. Alternatively, other arrangements may be possible, such asremotely locating processing unit 104 from receiver 102.

[0037] According to the disclosed embodiments, radar system 10 mayprovide the mechanism to obtain radar parameters, such as ground movingtarget indication (“GMTI”), air moving target indication (“AMTI”), andsynthetic aperture radar (“SAR”) imaging by forming simultaneoustransmitter and receiver beams. The gain and directivity of each of thesimultaneous transmitter and receiver beams may be comparable to knownsystems, such as phased array radar and bistatic radar technology. Usingradar system 10, the transmitter and receiver beamforming may becontrolled by a user within the field of view of the transmitter,provided the user has knowledge of the radar waveform codes. Thus, radarfunctions may be provided on-demand over wide geographical areas. Theradar transmitter may be shared by multiple users over a widegeographical area without the need for specific requirements to task thesource of illumination. For example, referring to FIG. 1, transmitter111 and receiver 102 are not coupled to each so as to exchange data.

[0038] The transmitter and receiver beams may be formed and adaptedafter the radar signals have been digitized by receiver 102 withinreceiving system 100. For AMTI and GMTI functions, displaced phasecenter aperture (“DCPA”) and space-time adaptive processing (“STAP”)algorithms may be applied to transmitter and receiver degrees offreedom. STAP algorithms may help mitigate interference and clutterproblems within receiving system 100. The availability of degrees offreedom that are physically located at the transmitter may be used tomotion compensate independently the transmitter and receiver in air andspace based systems. As a result, clutter may be suppressed in anairborne or space based system with as few as four degrees of freedom.The distribution of degrees of freedom between the transmitter andreceiver may be used to extend other clutter suppression techniques,such as doppler nulling, from monostatic and bistatic systems. Dopplernulling may be defined as eliminating spatially and spectrallyconcentrated noise. For SAR functions, users may form single transmitterbeams that remain focused on a user-specified center for spotlightimaging. Alternatively, transmitter beams may be formed to support scanmode imaging.

[0039] STAP performance may be dependent on scattered interference,available degrees of freedom, available processing power, and cost. STAPoperations may take advantage of all the information available to radarsystem 10 to cancel interference adaptively. A STAP enabled radar system10 may be able to dynamically respond to changes in the interferenceenvironment. Independent channels in space or time may be referred to asdegrees of freedom. A STAP enabled system collects information from theindependent channels and may use the information to compute the optimumweighting to accomplish the goal. STAP processing may minimize, orcancel, clutter and jamming while preserving the target signal in adesired direction. A reduction of minimum detectable velocity may bepossible using STAP.

[0040] The primary degrees of freedom are in space and time. Spatialdegrees of freedom are provided by the outputs of the array elements ofradar system 10. Time degrees of freedom are formed by delayed replicasof the outputs from the array elements, or time taps. Other potentialSTAP degrees of freedom may include beam outputs and Doppler filteroutputs (for post-Doppler or Beamspace STAP). Processing requirementsare also an important factor in the cost of STAP. Too many adaptivedegrees of freedom may overwhelm typical radar system signal processors.Transmitter degrees of freedom also may depend on the number ofelements, or sub-apertures, associated with the transmitter array.

[0041] Thus, the disclosed embodiments may use “noise-like” waveformsfor radar to extend the coverage area. Further, the disclosedembodiments may distribute spatially the radar degrees of freedom toobtain an increased level of clutter and interference suppression.Alternatively, the disclosed embodiments may enable clutter suppressionwith a smaller number of degrees of freedom than known systems. Clutterand interference suppression may be achieved by using orthogonal andpseudo-orthogonal transmitted waveforms to provide a dual transmit andreceive aperture adaptivity. This feature may enable a reduction in therequired sensor system degrees of freedom.

[0042] According to the disclosed embodiments, an adaptive broadcastradar system is disclosed that reduces the degrees of freedom within thesystem and enables the spatial distribution of degrees of freedombetween a transmitter and a receiver. For example, the disclosedembodiments may be used to suppress clutter in an airborne orspace-based system with as few as four degrees of freedom. Similarly,these degrees of freedom may be distributed spatially between thetransmitter and the receiver according to the prevailing circumstances.A more robust transmitter may be implemented in conjunction with a lessrobust set of receivers. This configuration may be desired for thosesystems adapted to a surveillance environment.

[0043] Radar system 10 may share transmitter resources among a widevariety of users such that a priori tasking or the control oftransmitter resources is not critical. Additional processing,intelligence, or tasking is avoided at the transmitter, other thanoperations to code independent waveforms. Radar functions may beperformed against objects, or targets, anywhere within the field-of-viewof the independently coded sub-arrays within the transmitter. Wherecoding occurs at the element level, the radar functions may be performedagainst objects anywhere in the forward hemisphere of the transmitterarray by any user with suitable receiving equipment and knowledge of thetransmitter waveform codes. The transmitted signal waveforms may bereconstructed at the receiever, thus, possibly eliminating the need fora directed transmitted beam.

[0044] The transmitter waveforms are regenerated at the receiver andeach user may form all potential beams independently, or any subset ofpotential transmitter beams. Thus, the transmitter waveforms arerecreated at the receiver without being coupled directly to thetransmitter. The transmitter beam pattern may be adapted by a user forpurposes that include spotlight or scan mode SAR imaging, andinterference/clutter suppression. Adaptive processing may include motioncompensation of the transmitter. When combined with the motioncompensation of the receiver, adaptive processing may provide bistaticdisplaced phase center aperture (“DPCA”) clutter suppression.

[0045] The transmitter antenna phase centers may be added to the degreesof freedom for STAP algorithms. Data for STAP processing is organizedinto 3-dimensional arrays, where the 3-dimensional index identifies thetransmitter/receiver array element number, the relative delay anddoppler associated with a batch or coherent processing interval of data,and the time associated with the batch or coherent processing intervalof data. The three-dimensional array may be known as data cubes. Theinclusion of additional degrees of freedom associated with thetransmitter array transforms a data cube into a new data structuretermed a data quad. The STAP formulation enables a large class ofemerging STAP techniques and algorithms to be developed for monostaticradars to be directly reformulated for bistatic radar systems, such asradar system 10.

[0046]FIG. 2 depicts a block diagram of a transmitter for an adaptivebroadcast radar system in accordance with an embodiment of the presentinvention. Transmitter 200 may be used in the adaptive broadcast radarsystem disclosed above. Transmitter 200 includes sub-apertures 210, 220,and 230. Transmitter 200 may have N number of sub-apertures, and is notlimited to the number disclosed with reference to FIG. 2. Sub-apertures210, 220, and 230 include waveform generators 212, 222, and 232,respectively. Waveform generators 212, 222, and 232 produce independentpseudo-random phase samples. Clock 202 is coupled to wave formgenerators 212, 222, and 232.

[0047] In addition to waveform generator 212, sub-aperture 210 includestime aperture 214 and amplifier 216. Sub-aperture 210 also includesphase shifters 218 and sub-aperture weights 219. In addition to waveformgenerator 222, sub- aperture 220 includes time aperture 224 andamplifier 226. Sub-aperture 220 also includes phase shifters 228 andsub-aperture weights 229. In addition to waveform generator 232,sub-aperture 230 includes time aperture 234 and amplifier 236.Sub-aperture 232 also includes phase shifters 238 and sub-apertureweights 239. Sub-aperture 230 may be the Nth aperture within transmitter200.

[0048] Transmitter 200 preferably is a phased array antenna withmultiple phase centers, and with independent signals transmitted on eachelement, or sub- aperture, of the array. The set of transmitted signalsmay be selected to orthogonal or pseudo-orthogonal. Orthogonal orpseudo-orthogonal signals may be generated using waveform coding, suchas Gold codes. The signal spectrum may be shared between transmitelements, or sub-apertures, using code division multiple access(“CDMA”). Alternatively, the transmit signals may be assigned toindependent frequency channels, and the spectrum shared using frequencydivision multiple access (“FDMA”). FDMA approaches also may includeguard bands within the signal spectrum to accommodate anticipateddoppler shifted clutter and targets. Thus, near orthogonality may becompleted. Polarization state also may be used to define orthogonaltransmit channels.

[0049] Thus, sub-aperture 210 tasks waveform generator 212 to generate awaveform from independent pseudo-random phase samples. The waveform ispassed to time aperture 214. Time apertures may create a train of pulsesthat are coded differently for each sub-aperture. Alternatively, timeaperture 214 may be bypassed to provide continuous wave (“CW”) codedsignals. For example, pseudo-random phase coding may occur where thelength of the sequence exceeds the expected coherent correlationintervals. The waveform or train of pulses is received by amplifier 216and passed to a bank of phase shifters 218 and sub-aperture weights 219.Phase shifters 218 and sub-aperture weights 219 may be set independentlyfor each sub-aperture. Phase shifters 218 and sub-aperture 219 may bedesigned to generate a fixed beam in a direction relative to theboresight of transmitter 200. Further, the waveform is coded with theinformation to generate the data quads at the receiver, as disclosed ingreater detail below. The waveform may be forwarded to antennas 280 fortransmission. Sub-apertures 220 and 230 may operate in a similar manner,such that different waveforms are transmitted by antennas 280.

[0050] The formation of sub-apertures allows the number of degrees offreedom to be limited, and, thus, reduce system complexity. Acorresponding reduction in coverage may occur where the coverage area isdefined by the sub-aperture beam patterns. An alternative approach toreduce system complexity may involve operations at lower frequencies.The use of lower frequencies may reduce the number of transmitterelements desired for full coverage in the forward hemisphere oftransmitter 200.

[0051]FIG. 3 depicts a block diagram of a receiver for an adaptivebroadcast radar in accordance with an embodiment of the presentinvention. Receiver 300 may receive transmitted signals directly orindirectly from transmitter 200 with antennas 380. Receiver 300 includesa clock 302 and sub-apertures 310, 320, and 330. Receiver 300 may have Nnumber of sub-apertures, and is not limited to the number disclosed withreference to FIG. 3. Like transmitter 200, receiver 300 employs banks ofphase shifters and sub-aperture weights within sub-apertures 310, 320,and 330 to modify the received signals from antennas 380. The formationof the sub-apertures may limit the number of degrees of freedom toreduce system complexity.

[0052] Sub-apertures 310, 320, and 330 also include receivers 316, 326,and 336, respectively. Receivers 316, 326, and 336 may be low-noise,high dynamic range receivers. In receivers 316, 326 and 336,sub-aperture formation may be used to reduce cost and complexity of theradar system if the adaptive broadcast radar transmitter, such astransmitter 200, is utilized for surveillance or reconnaissance in arestricted area. To support operations that desire wide area coverage,the radar system may operate at a lower frequency to reduce systemcomplexity without a corresponding reduction in coverage. Receivers 316,326, and 336 are coupled to adaptive broadcast radar transmitter channeldemultiplexers 350, 360, and 370, respectively. Preferably,demultiplexer is dedicated to each sub-aperture.

[0053] The sampled output of the k^(th) sub-aperture in the receiversystem may be denoted V_(Rx) (k, t_(n)). Motion compensation functions318, 328, and 338 may remove time dependent phase delays between thetransmitter and receiver system. Motion compensation may be performedfor each receiver sub-aperture independently. Because the receivedsignal is a composite of transmitted signals, a single point on thetransmitter, such as transmitter 200, may be motion compensated. Thetransmitter may be known as the j₀ ^(th) transmitter sub-aperture. Tosimplify the derivations, a scaling also may be included in motioncompensation functions 318, 328, and 338. The scale factor may be theinverse of the transmitted signal strength. The signal strength may beproportional to the square-root of the transmitter power delivered tothe j₀ ^(th) transmitter sub-aperture. The motion compensated signal maybe given by:${X_{k}(t)} \equiv {\frac{V_{RX}\left( {k,t} \right)}{V_{Tx}(t)}^{{- 2}\quad \pi \quad \quad {f{({t - {\tau {({{{\overset{\_}{X}}_{{Tx}:{j0}}{(t)}} - {{\overset{\_}{X}}_{{RX}:k}{(t)}}})}}})}}}}$

[0054] where f is the center frequency of the transmitted signal, and,τ({overscore (X)}_(Tx:j0)(t)−{overscore (X)}_(Rx:k)(t)) is the signalpropagation delay from the j₀ ^(th) transmitter sub-aperture to thek^(th) receiver sub-aperture. For example, transmitter sub-aperture 210may send a signal to receiver sub-aperture 310 that is motioncompensated. The above algorithm discloses the motion compensationoperation performed by motion compensation function 318.

[0055] Receiver 300 also includes signal processor 390. Signal processor390 may be coupled to the sub-apertures of receiver 300, such assub-apertures 310, 320, and 330. Signal processor 390 may correlate toprocessing unit 104 depicted in FIG. 1. Signal processor 390 forms thedata quads for STAP operations within the adaptive broadcast radarsystem. Signal processor 390 is disclosed in greater detail below. Usingthe respective demultiplexers and signal processor 390, the transmittedsignal waveform may be regenerated from the information encoded onto thewaveform at transmitter 200. Receiver 300, as disclosed in greaterdetail below, is capable of recreating the transmit signal beam from thenumerical data encoded on the waveform, such as degrees of freedom oftransmitter 200 without being directly coupled to transmitter 200.

[0056]FIG. 4 depicts a block diagram of an adaptive broadcast radarchannel demultiplexer in accordance with an embodiment of the presentinvention. Demultiplexer 400 correlates to demultiplexers 350, 360, and370 depicted in FIG. 3. Demultiplexers 350, 360, and 370, however, arenot limited to the embodiments disclosed by FIG. 4. Demultiplexer 400includes a bank of waveform generators 402, 404, and 406. Waveformgenerators 402, 404, and 406 are time-synchronized with the transmitter,such as transmitter 200, and may generate replicas of each of the Ntransmitted signals, where N is the number of transmitter sub-apertures.Thus, demultiplexer 400 may have N waveform generators correlating to Nsub-apertures of transmitter 200. Waveform generators 402, 404, and 406are coupled to clock 302.

[0057] Waveform compensation may be initialized and/or updated once percoherent processing interval. The coherent processing interval is chosensuch that the number of signal samples is greater than or equal to J, #or the number of transmitter sub-apertures, times M, or the number ofdelay values desired to cover the ground clutter grid. A waveformcompensation filter computation function 408 may generate and format anN_(t) x (J·M) array of delayed reference signals, where N_(t) may be thenumber of samples in a coherent processing interval. The referencesignal data, s_(j) (t_(n)−τ_(m)), associated with the m^(th) delay forthe j^(th) transmitter sub-aperture is mapped into the q^(th) column,where q(j,μ)=μ+(j−1)·M. The inverse map of the generalized index, q,into the sub-aperture index, j, and the delay index, μ, may be given by:

μ≡mod(q,M) $j \equiv {{floor}\quad \left( \frac{q}{M} \right)}$

[0058] For the coherent processing interval starting with the timesample t_(n0), the array of delay-compensated reference data may begiven by:

Σn,q(n ₀)≡s _(j(q))(t _(n)−τ_(μ(q)))

[0059] The term also may be written in terms of {tilde over(S)}_(j, n)≡S_(j)(t_(n)). Because t_(n)−τ_(μ(q))=t_(n−μ(q)) andS_(j)(t_(n)−τ_(μ))={tilde over (S)}j,n−m, the array of delay-compensatedreference may be given by: ${\sum({n0})} \equiv \begin{pmatrix}{{\overset{\sim}{S}}_{0},_{n_{0}}} & {{\overset{\sim}{S}}_{0},_{n_{0} - 1}} & \ldots & {{\overset{\sim}{S}}_{0},_{n_{0} - M + 1}} &  & {{\overset{\sim}{S}}_{1},_{n_{0}}} & {{\overset{\sim}{S}}_{1},_{n_{0} - 1}} & \ldots & {{\overset{\sim}{S}}_{0},_{n_{0} - M + 1}} & {\ldots } & {{\overset{\sim}{S}}_{J},_{n_{0}}} & {{\overset{\sim}{S}J},_{n_{0} - 1}} & \ldots & {{\overset{\sim}{S}J},_{n_{0} - M + 1}} \\{{\overset{\sim}{S}}_{0},_{n_{0} - 1}} & {{\overset{\sim}{S}}_{0},_{n_{0} - 2}} & \ldots & {{\overset{\sim}{S}}_{0},_{n_{0} - M}} &  & {{\overset{\sim}{S}}_{0},_{n_{0} - 1}} & {{\overset{\sim}{S}}_{1},_{n_{0}}} & \ldots & {{\overset{\sim}{S}}_{0},_{n_{0} - M}} & {\ldots } & {{\overset{\sim}{S}J},_{n_{0} - 1}} & {{\overset{\sim}{S}J},_{n_{0}}} & \ldots & {{\overset{\sim}{S}J},_{n_{0} - M}} \\\ldots & \ldots & \ldots & \ldots &  & \ldots & \ldots & \ldots & \ldots & {\ldots } & \ldots & \ldots & \ldots & \ldots \\{{\overset{\sim}{S}}_{0},_{n_{0} + {Nt} - 1}} & {{\overset{\sim}{S}}_{0},_{n_{0} + {Nt} - 2}} & \ldots & {{\overset{\sim}{S}}_{0},_{n_{0} + {Nt} - M}} &  & {{\overset{\sim}{S}}_{1},_{n_{0} + {Nt} - 1}} & {{\overset{\sim}{S}}_{1},_{n_{0} + {Nt} - 2}} & \ldots & {{\overset{\sim}{S}}_{0},_{n_{0} + {Nt} - M}} & {\ldots } & {{\overset{\sim}{S}J},_{n_{0} + {Nt} - 1}} & {{\overset{\sim}{S}J},_{n_{0} + {Nt} - 2}} & \ldots & {{\overset{\sim}{S}J},_{n_{0} + {Nt} - M}}\end{pmatrix}$

[0060] If N_(t) is selected such that N_(t)=J·M, summation Σ may be asquare matrix, and, for pseudo-random phase code signals, it may beshown to be invertible. Accordingly, waveform compensation filtercomputation function 408 may be Σ⁻¹. Thus, when N_(t)>J·M, thepseudo-inverse may be used and waveform compensation filter 408 may begiven by:

(Σ*^(T)Σ)⁻¹Σ*^(T)

[0061] The output of waveform compensation filter 408, also known as thechannel transfer function, is given by H=W^(T)Y.

[0062] When the channel transfer functions H include both delayed anddoppler-shifted signal components, summation Σ may be replaced by anarray of reference signals that are compensated for both delay, τ_(μ),and doppler shift, f_(v). A generalized index q defines the column fordata associated with the j^(th) sub-aperture, the μ^(th) delay andv^(th) doppler and

q(j,μ,v)≡v+[μ+(j−1)·M]·N

[0063] The inverse map, of the generalized index, q, into thesub-aperture index, j, and delay index, μ, may be given

v≡mod(q−1, N)+1$\mu \equiv {{{mod}\quad \left( {{{floor}\quad \left( \frac{q - 1}{N} \right)},M} \right)} + 1}$$j \equiv {{{floor}\quad \left( \frac{q - 1}{M \cdot N} \right)} + 1}$

[0064] Then, the array of compensated reference signals may be given by:${\sum_{n,q}\left( n_{0} \right)} \equiv {^{2\quad \pi \quad \quad {{fv}{(1)}}{({{tn} - \tau_{\mu {(q)}}})}_{S_{j{(q)}}}}\left( {{tn} - \tau_{\mu {(q)}} - {\frac{\lambda \quad f_{v{(q)}}}{c_{light}}\left( {t_{n} - \tau_{\mu {(q)}}} \right)}} \right)}$

[0065] where n ε[n₀, n₀+N_(t)−1] and q ε [1, J·M·N−1].

[0066] For the k^(th) receiver system sub-aperture, H is a vector oflength J·M·N. Vector H may be reformatted into a J×(M×N) array where thej^(th) element discloses the dependence of the channel transfer functionon transmitter sub-array degrees of freedom and (μ,v) discloses thedelay and doppler dependence.

[0067] Thus, according to the disclosed embodiments, demultiplexer 400receives the motion compensated signals Y_(n) at input data formatfunction 412. Input data format function 412 also receivessynchronization data from clock 302. Waveform generators 402, 404, and406 also are synchronized by clock 302 to generate replicas of thetransmitted signals. These replica signals are passed to waveformcompensation filter function 408, also synchronized with clock 302.Waveform compensation filter function 408 outputs compensated referencesignals to channel transfer function 410, which outputs channel transferfunctions 490.

[0068] Channel transfer functions 490 contain delay, doppler, and otherinformation about the received signal data. Channel transfer functions490 may be specific to received signal data from each sub-aperture ofreceiver 300.

[0069]FIG. 5 depicts a block diagram of a signal processor in accordancewith an embodiment of the present invention. Signal processor 500enables the formation of data quads for STAP applications. Because ofthe simultaneous transmitter and receiver beams, the steering vector, G,is a quad that discloses the desired sensor response for bothtransmitter 200 and receiver 300. STAP may be a framework rather than aspecific algorithm. The definition of the steering vector and thedefinition of the approach to modeling and estimating the measurementcovariance are desired to transform the STAP framework into a specificalgorithm.

[0070] The STAP covariance may be modeled as a diagonal matrix,independently of delay and time to provide simultaneous fixedtransmitter and receiver beamforming. Weight vectors are proportional tothe steering vector and provide the amplitude and phase adjustmentsdesired to steer the transmitter and receiver beams in a particulardirection. Diverse types of fully adaptive and partially adaptive STAPalgorithms designed to adapt both transmitter and receiver antennapatterns may be specified in terms of the weight vectors. For example,bistatic DPCA may be implemented using steering vector components may bedefined by: $G = \begin{Bmatrix}{g_{j} \otimes g_{k}} \\{{- 2}{g_{j} \otimes g_{k}}} \\{g_{j} \otimes g_{k}}\end{Bmatrix}$

[0071] Thus, signal processor 500 may operate as follows. Channeltransfer functions 490 are received from demultiplexer 400 at channeltransfer data function 512. Channel transfer data function 512 formatsthe channel transfer functions 490 into channel transfer components, orH={h_(j,k) (t_(n m))}. Steering vector computation function 510 receivescoherent processing interval parameters from clock 302 and computessteering vector components, or G={g_(j,k) (t_(n m))}. STAP data quadreformat functions 516 and 514 then reformat the channel transfercomponents and steering vector components, respectively.

[0072] STAP covariance function 518 receives the channel transfercomponent H and computes, or estimates, a measurement covariance. TheSTAP covariance may be depicted as a diagonal matrix, and independent ofdelay and time to provide simultaneous fixed transmitter and receiverbeamforming. STAP weight vector function 520 receives the measurementcovariance and the steering vector component, G, to compute the weightvector, or W=R⁻¹G. Weight vectors may be proportional to the steeringvector, G. Weight vectors may provide the amplitude and phaseadjustments desired to steer the transmitter and receiver beams in aparticular direction. This feature allows diverse types of fullyadaptive and partially adaptive STAP algorithms for both transmitter andreceiver antenna patterns to be specified in terms of the weightvectors.

[0073] The weight vectors are received at filtered data function 522,along with channel transfer components, or H. The channel data withinthe channel transfer components may be reformatted and the data quadscomputed by applying the weight vectors to the channel transfercomponents. Thus, data quad 530 is output to other system components forfurther processing. Data quad 530 may be defined in four-dimensions byindices determined by filtered data function 522.

[0074]FIG. 6 depicts a flowchart for formatting data within the receivedsignal in accordance with an embodiment of the present invention. Thedata is formated to form the data quads disclosed above. The signal maybe received at receiver 300, and the following disclosed operations maybe executed within receiver 300, demultiplexer 400, and signal processor500. Step 602 executes by compensating the receiver data from thesignal. Specifically, the receiver data is motion compensated byprocessing the data associated with each element, or sub- aperture, ofthe receive array of receiver 300. Step 604 executes by removing thedoppler shift from the receiver data. Step 604 may be executed inconjunction with step 602. The doppler shift may exist due to therelative motion of receiver 300 and transmitter 200. The doppler shiftremoval may be given by

X _(k)(t)=X ₀ _(k) (t)e ^(−2πif) ^(_(Tx−Rx)) ^(t)

[0075] Step 606 executes by estimating the direct path component, ortransmitter signal, of the received signal. The total transmitted signalmay be modeled parametrically and the model parameters estimated. Thetransmitter signal may be modeled on an element-by-element basis and themodel may include parameters to describe the bearing of receiver 300relative to the boresight of transmitter 200, and the magnitude of thetransmitted signal power at the element of receiver 300. Modelparameters may be estimated using known adaptive techniques to minimizereceived power. Step 608 executes by cancelling the direct pathcomponent, or transmitter signal from the received signal. Specifically,estimates of the direct path signal, as disclosed above, may besubtracted from the received data, given by${Y_{k}(t)} = {{X_{k}(t)} - {a_{k}{\sum\limits_{j = 1}^{j}{\left\{ ^{2\pi \quad {{({j - 1 - \frac{j - 1}{2}})}}\delta_{L}{\sin({(\phi_{k})}}} \right\} S_{j}}}}}$

[0076] Step 610 executes by segmenting the received data into thecoherent processing intervals. Coherent processing intervals also may beknown as “dwells.”W=R⁻¹G. As disclosed with reference to FIG. 5, theweight vectors provide the amplitude and phase adjustments desired tosteer beams in a particular direction.

[0077] Step 622 executes by reformatting the data quad, and itsinformation, with the weight vector. The data quad includes informationon the transmitter/receiver element number, the delay and/or dopplerassociated with the batch of data for the coherent processing interval,the time for the coherent processing interval, and the degrees offreedom. The information in the data quad describes the transmittedsignal such that it may be regenerated by receiver 300. Step 624executes by forwarding the data quad information to additionalprocessing components.

[0078] The following examples may be applied to bistatic radar. First,an extension of DPCA may be considered. The extension may be designed toprovide adaptation in the transmit and receive arrays.

[0079] The equation Let$g_{Tx} = \left\{ ^{2\quad \pi \quad {{({k\frac{K + 1}{2}})}}\delta \quad {\sin {(\phi_{Tx})}}} \right\}$

[0080] may represent the transmit aperture weights designed to provide afixed beam in direction, φ_(Tx), relative to the boresight oftransmitter 200. In addition, the equation let$g_{Rx} = \left\{ ^{2\quad \pi \quad {{({k\frac{K + 1}{2}})}}\delta \quad {\sin {(\phi_{Rx})}}} \right\}$

[0081] may define the desired weights for the receive system as depictedwith reference to FIG. 3. W=R⁻¹G. As disclosed with reference to FIG. 5,the weight vectors provide the amplitude and phase adjustments desiredto steer beams in a particular direction.

[0082] Step 622 executes by reformatting the data quad, and itsinformation, with the weight vector. The data quad includes informationon the transmitter/receiver element number, the delay and/or dopplerassociated with the batch of data for the coherent processing interval,the time for the coherent processing interval, and the degrees offreedom. The information in the data quad describes the transmittedsignal such that it may be regenerated by receiver 300. Step 624executes by forwarding the data quad information to additionalprocessing components.

[0083] The following examples may be applied to bistatic radar. First,an extension of DPCA may be considered. The extension may be designed toprovide adaptation in the transmit and receive arrays.

[0084] The equation Let$g_{Tx} = \left\{ ^{2\quad \pi \quad {{({k\frac{K + 1}{2}})}}\delta \quad {\sin {(\phi_{Tx})}}} \right\}$

[0085] may represent the transmit aperture weights designed to provide afixed beam in direction, φ_(Tx), relative to the boresight oftransmitter 200. In addition, the equation let$g_{Rx} = \left\{ ^{2\pi \quad {{({k\frac{K + 1}{2}})}}\delta \quad {\sin {({\phi \quad {Rx}})}}} \right\}$

[0086] may define the desired weights for the receive system as depictedwith reference to FIG. 3. G may be formed as follows:$G = {\left\{ {{- 2}\begin{matrix}{g_{Tx} \otimes \quad g_{Rx}} \\{g_{Tx} \otimes \quad g_{Rx}} \\{g_{Tx} \otimes \quad g_{Rx}}\end{matrix}} \right\}.}$

[0087] G may correspond to a three-pulse DPCA. Second, G may be definedas the steering vector for a target doppler, f_(d), or$G = \left\{ {- \begin{matrix}{{g_{Tx} \otimes \quad g_{Rx}}^{{- 2}\pi \quad \quad f_{d}T}} \\{g_{Tx} \otimes \quad g_{Rx}} \\{{g_{Tx} \otimes \quad g_{Rx}}^{2\pi \quad \quad f_{d}T}}\end{matrix}} \right\}$

[0088]FIG. 7 depicts a flowchart for generating a sensor signal model inaccordance with an embodiment of the present invention. A sensor signalmodel may facilitate removing the pseudo-random baseband modulation fromthe received signal. The sensor signal model may disclose the receivedsignal in terms of the clutter and target environemnt, and is desired tointerpret effectively the sensor output. Step 702 executes by defining aclutter component for the sensor signal model. According to thedisclosed embodiments,${X_{p}\left( {k,t} \right)} = {{\sum\limits_{j = 0}^{J - 1}\left\{ {{{A_{{0:j},k}\left( {{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}} \right)}^{{\psi}_{jk}{(t)}}} + {\int\limits_{A_{clutter}}{{A_{{c:j},k}\left( {{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}} \right)}^{{- }\quad {\varphi_{c\quad {jk}}{({{\overset{\_}{x}}_{c},t})}}}^{{\psi}_{jk}{({t - {\Delta \quad {\tau_{cjk}{({{\overset{\_}{x}}_{c},t})}}}})}}{^{2}{\overset{\_}{x}}_{c}}}}} \right\}} + {v(t)}}$

[0089] where Φ^(j) is the phase of the baseband signal transmittedthough the jth sub-aperture. Ψ^(j) _(k) may be equal to a delayedversion of Φ^(j) where the delay may be about equal to the direct pathdelay from the reference transmitter sub-aperture, j₀, to the kthreceiver sub-aperture, or

Ψ_(k)(t)≡Φ^(j)(t−τ({overscore (x)} _(Txj) ₀ (t)−{overscore (x)}_(Rx:k)(t)))

[0090] Δτ_(c:jk) (x_(c),t) may be the delay from transmitter 200 toclutter patch to receiver 300 relative to the direct path delay asdisclosed above. In particular, the delay may be given by

Δτ_(c:jk)(x _(c) ,t)≡τ(R _(jc)(t))+τ(R _(ck)(t))−τ(R _(0:j) ₀ _(k)(t))

[0091] φ_(c:jk) (x_(c),t) may be the delay from transmitter 200 toclutter patch to receiver 300 relative to the direct path phase delay.The RF phase delay may be given by

φ_(c:jk) ({overscore (x)} _(c) , t)≡2πf ₀Δτ_(c:jk)({overscore (x)} _(c), t)

[0092] A_(0:j,k) may be the relative strength of the direct path signalrelative to the transmitted signal strength. G_(Tx:jk) may denote thesub-aperture gain of the j^(th) transmitter sub-aperture in thedirection of the k^(th) receiver sub-aperture. G_(Rx:jk) may denote thesub-aperture gain of the k^(th) receiver sub-aperture in the directionof the j^(th) transmitter sub-aperture. L may denote the path loss forthe direct path signal component. The relative strength of the directpath signal relative to the transmitted signal strength may be given by

A _(0;j,k)({overscore (x)} _(Tx) ,{overscore (x)} _(Rx))≡└G _(Tx:jk) G_(Rx:jk) L(R _(0:jk))┘

[0093] A_(c:j,k) may be the relative strength of the scattered signalwhere {overscore (x)}_(c) may denote the location of the clutter patch,ê_(jc) and ê_(kc) may be unit vectors that point from transmitter 200 tothe clutter patch and from the clutter patch to receiver 300.σ⁰({overscore (x)}_(c)|ê_(jc),ê_(ck)) may be the relative bistaticreflectivity of the ground patch at {overscore (x)}_(c). Thus, therelative strength of the scattered signal may be given by

A _(c:j,k)({overscore (x)} _(c) ,{overscore (x)} _(Tx) ,{overscore (x)}_(Rx))≡└G _(Tx)(ê _(jc)) G _(Rx)(ê _(ck))L(R _(jc))L(R_(ck))σ⁰({overscore (x)} _(c) |ê _(jc) , ê _(ck))┘

[0094] In addition v(t) may denote the noise process.

[0095] Step 704 executes by generating integral of the received signal.The received signal may be re-expressed in terms of a continuousintegral over the delay measurement, τ, or${X_{p}\left( {k,t} \right)} = {{\sum\limits_{j = 0}^{J - 1}\left\{ {{{H_{{0:j},k}(t)}{V_{jk}(t)}} + {\int\limits_{\Delta \quad \tau}{{H_{{c:j},k}\left( {\tau^{\prime},t} \right)}{V_{jk}\left( {t - \tau^{\prime}} \right)}{\tau^{\prime}}}}} \right\}} + {v(t)}}$

[0096] where V_(jk) (t)≡e^(iΨ) ^(j) ^(k(t)).

[0097] Step 706 executes by defining the bistatic channel transferfunction (“BCTF”). H_(c:j,k) (τ,t) may be termed the BCTF and may begiven by an integral over a constant delay strip, or${H_{{c:j},k}\left( {\tau,t} \right)} \equiv {\int\limits_{\lbrack{{\Delta \quad {\tau_{c}{({\overset{\_}{x}}_{c})}}} = \tau}\rbrack}{^{{- }\quad {\phi_{c\quad {jk}}{({{\overset{\_}{x}}_{c},t})}}}{A_{{c:j},k}\left( {{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}} \right)}{^{2}{\overset{\_}{x}}_{c}}}}$

[0098] Step 708 executes by linearizing the phase delay of the BCTF.Linearization of the phase delay in the BCTF may demonstrate thedependence on doppler and the bearing of transmitter 200 and receiver300 to the clutter patch, or

φ_(c:jk)({overscore (x)} _(c) ,t)=φ₀ +K ^(T) ·D _(p)

[0099] where $D_{p} = \begin{bmatrix}{{\,^{d}{Tx}}\left\lfloor {{\sin \left( {\,^{\phi}{Tx\_ c}} \right)} + {\sin \left( {\,^{\phi}{Tx\_ Rx}} \right)}} \right\rfloor} \\{{\,^{d}{Rx}}\left\lbrack {{\sin \left( {{\,^{\varphi}{c\_ Rx}} + {\,^{\eta}{Rx}}} \right)} - {\sin \left( {{\,^{\varphi}{Tx\_ Rx}} + {\,^{\eta}{Rx}}} \right)}} \right\rbrack} \\{{{{\,^{V}{Tx}}\left\lbrack {{\sin \left( {\,^{\varphi}{Tx\_ c}} \right)} + {\sin \left( {\,^{\varphi}{Tx\_ Rx}} \right)}} \right\rbrack}\delta \quad t_{nyquist}} + {{\,^{V}{Rx}}\left\lbrack {{\sin \left( {{\,^{\varphi}c} + {Rx}} \right)} -} \right.}} \\{\left. {\sin \left( {\,^{\varphi}{Tx\_ Rx}} \right)} \right\rbrack \delta \quad t_{nyquist}}\end{bmatrix}$

[0100] Step 710 executes by absorbing the constant phase term into therelative strength of the scattered signal, or A_(c:j,k). Thus, the BTCFmay be given by${H_{{c:j},k}\left( {\tau,t} \right)} \equiv {\int\limits_{\lbrack{{{\Delta\tau}_{c}{({\overset{\_}{x}}_{c})}} = \tau}\rbrack}{^{{- }\quad K^{T \circ_{D_{P}}}}A_{{c:j},{k{({{\overset{\_}{x}}_{c},{\overset{\_}{x}}_{Tx},{\overset{\_}{x}}_{Rx}})}}}{_{{\overset{\_}{x}}_{c}}^{2}.}}}$

[0101] Step 712 executes by defining the channel transfer function. Thesignal may be sampled at the in-time at the Nyquist rate and in delay ata rate consistent with the delay resolution, δτ. Step 714 executes bygenerating a sampled version of the received signal. The sampled versionof the received signal may be a function of the channel transferfunction, or given by${X_{p}\left( {k,n} \right)} = {{\sum\limits_{j = 0}^{J - 1}{\sum\limits_{m = 0}^{M - 1}{{H_{{c:j},k}\left( {m\quad {\delta\tau}} \right)}{V_{jk}\left( {{n\quad \delta \quad t_{nyquist}} - {m\quad \delta \quad \tau}} \right)}\delta \quad \tau}}} + {v_{k,n}.}}$

[0102] Step 716 executes by revising the signal model. The focus of thisstep is on associating with the k₀ ^(th) receiver sub-aperture. Usingsimplified notation and dropping the indices associated with thesub-aperture of receiver 300 and the coherent processing interval, thesignal model may be rewritten to$Y_{n} = {{\sum\limits_{j = 0}^{J - 1}{\sum\limits_{m = 0}^{M - 1}{H_{j,m}S_{j,{n - m}}}}} + v_{n}}$

[0103] where

[0104] Y_(n)≡X_(p) (k,n)

[0105] H_(j,m)≡H_(c:j,k) (mδτ)δτ

[0106] S_(j,n)≡V_(jk) (nδτ_(nyquist))

[0107] Thus, the signal model for a batch of data associated with the k₀^(th) receiver sub-aperture and the time sample n=n₀ ma be given by

Y≡{X _(k) ₀ _(,n)}_(nε[n) ₀ _(,n) ₀ _(+N−1])

[0108] Step 718 executes by generating a linear system model for thesignal model. The linear system model may be expressed as$\begin{pmatrix}{Y(0)} \\{Y(1)} \\ \\{Y\left( {N - 1} \right)}\end{pmatrix} = {\begin{pmatrix}H_{0,0} \\H_{0,1} \\ \\H_{0,{M - 1}} \\H_{1,0} \\H_{1,1} \\ \\H_{1,{M - 1}} \\\underset{\_}{\cdots} \\H_{{J - 1},0} \\H_{{J - 1},1} \\ \\\underset{\_}{H_{{J - 1},{M - 1}}}\end{pmatrix} + \begin{pmatrix}{v(0)} \\{v(1)} \\ \\{v\left( {N - 1} \right)}\end{pmatrix}}$

[0109] The general lest squares solution may be give by {tilde over(H)}=(Σ^(T)Σ)⁻¹Σ^(T)Y where P≡Σ^(T)Y represents application of ageneralized matched filter to received data and R≡(Σ^(T)Σ) is awhitening, or waveform compensation filter, such as waveformcompensation filter 408. When N_(t)=J·M·N, Σ may be invertible and,thus, H=Σ⁻¹Y.

[0110] The disclosed process may remove the pseudo-random basebandmodulation from the received signal. The channel transfer function maystill have RF phase delay information encoded within. The phased delayinformation may be related to the direction of the clutter patchrelative to the specific transmitter and receiver sub-apertures. Theapplication of the steering vectors to the channel transfer functionsmay separate out the linear phase delays associated with specificdirections from transmitter 200 and receiver 300. Application oftransmitter aperture weights to the BCTF compensates for the linearphase term in the BCTF model. This action may provide a technique toresolve scattering sources in the delay and the angle of the receivedsignal.

[0111]FIG. 8 depicts a flowchart for performing radar operations withinan adaptive broadcast radar system in accordance with an embodiment ofthe present invention. FIG. 8 depicts the processes disclosed above,with reference to overall adaptive broadcast radar system. Within thesystem, a signal waveform is transmitted continuously from a transmitterover an area that may have a number of receivers. The receiver mayreceive signals from the transmitter and scattered signals reflected offa target within the area.

[0112] Step 802 executes by generating a signal waveform within thetransmitter, such as transmitter 200. The signal waveform may begenerated within a sub-aperture of transmitter 200. Preferably, eachsub-aperture of transmitter 200 may generate a signal in a pseudo-randommanner. Step 804 executes by encoding the signal waveform with numericalinformation. Specifically, the signal waveform is encoded orthogonally.By being encoded orthogonally, the signals from the differentsub-apertures may be distinguishable. Further, the degrees of freedomassociated with transmitter 200 also may be encoded into the signals.Step 806 executes by placing phase shifts and amplitude, or weights,onto the signal. The phase shifters and weights may be set independentlyfor each sub-aperture of transmitter 200.

[0113] Step 808 executes by transmitting the orthogonally encoded signalin a continuous manner over an area. Preferably, the area may be forwardof transmitter 200. The signals may be directed towards the area withthe purpose of reaching receivers within the area. Further, transmitter200 transmits from an array antenna coupled to the sub-apertures.Transmitter 200 may be in motion, such as an airplane pointingtransmitter 200 below itself to receivers on the ground. Step 810executes by receiving the direct and scattered signals at receiver 300.The received signal may be a composite of the transmitted signals, andmay have data for the transmitted signals. Receiver 300 also may have anarray antenna coupled to sub-apertures that correlate to thesub-apertures on transmitter 200. The signals received are digitallyreconstructed to determine target parameters from the direct pathsignals and scattered signals.

[0114] Step 812 executes by performing motion compensation to removetime dependent phase delays between transmitter 200 and receiver 300.Motion compensation may be performed independently for each sub-apertureof receiver 300. Step 814 executes by determining the delay and doppler,if applicable, for the received signal. This information may be placedinto a channel transfer function.

[0115] Step 818 executes by regenerating the transmit signal fromtransmitter 200. By using the information encoded in the transmittedsignal waveforms, such as degrees of freedom and the transmitsub-aperture or element number, and the information derived from thesignal, such as delay and doppler, the transmit beam may berecontructed. Although the transmitted signal was not “beamed” to aspecific location, receiver 300 may use the parameters identified aboveto “reconstruct” a beam transmitted at the target. The reconstructedbeam may be used for additional radar operations, such as targettracking. Step 820 executes by controlling the regenerated transmit beamat receiver 300 for tracking the target detected from the scatteredsignals.

[0116] It will be apparent to those skilled in the art that variousmodifications and variations can be made in the disclosed embodiments ofthe present invention without departing from the spirit or scope of theinvention. Thus, it is intended that the present invention embodies themodifications and variations of this invention provided that they comewithin the scope of the appended claims and their equivalents.

What is claimed is:
 1. A method for formatting received data within anadaptive broadcast radar system having a transmitter comprisingsub-apertures and a receiver comprising sub-apertures, wherein said datais received at said receiver, comprising: providing an estimate for adelay of scattered signal components within said received data;generating an index for said estimate, wherein said index includes atransmitter element number and a receiver element number; generating adata quad for said index; and estimating a measurement covariance and aweight vector for said data quad, wherein said data quad is reformattedwith said measurement covariance and said weight vector.
 2. The methodof claim 1, further comprising compensating said receiver data for amotion of said transmitter or said receiver.
 3. The method of claim 2,further comprising removing a doppler shift from said receiver data. 4.The method of claim 1, further comprising estimating a direct pathcomponent from said received data.
 5. The method of claim 4, furthercomprising cancelling said direct path signal from said received data.6. The method of claim 1, wherein said received data includes a codeencoded by said transmitter, said code including information about saidtransmitter.
 7. The method of claim 6, wherein said information includesdegrees of freedom.
 8. The method of claim 1, further comprisingsegmenting said received data according to a coherent processinginterval.
 9. The method of claim 8, wherein said segmented datacorrelates to said estimate.
 10. The method of claim 1, wherein saidestimate further includes a doppler delay.
 11. The method of claim 1,further comprising determining said weight vector from a steeringvector.
 12. The method of claim 1, wherein said estimating a measurementcovariance includes using a channel transfer function.
 13. A method forobtaining target parameters within an adaptive broadcast radar system,comprising: coding information about a signal waveform generated by atransmitter having sub-apertures; receiving a signal at a receiverhaving sub-apertures corresponding to said sub-apertures of saidtransmitter, wherein said received signal correlates to said signalwaveform; decoding information about said signal waveform from saidreceived signal; determining a data quad from said decoded information,wherein said data quad includes degrees of freedom associated with saidtransmitter.
 14. The method of claim 13, further comprising generatingsaid signal waveform within said sub-aperture of said transmitter. 15.The method of claim 13, further comprising applying a phase shift tosaid signal waveform within said transmitter sub-apertures.
 16. Themethod of claim 15, further comprising applying a weight vector to saidsignal waveform within said transmitter sub-apertures.
 17. The method ofclaim 16, further comprising motion compensating said received signal byremoving said weight vectors and said phase shifts.
 18. The method ofclaim 13, wherein said received signal is a composite of transmittedsignal from said signal waveform.
 19. The method of claim 13, furthercomprising generating a channel transfer function comprising delay anddoppler signal components of said received signal.
 20. The method ofclaim 19, wherein said determining includes formatting said channeltransfer function with a weight vector and measurement covariance ofsaid received signal.
 21. The method of claim 20, wherein said signalwaveform is transmitted as an orthogonal waveform.
 22. A method forgenerating a sensor signal model for a received signal within anadaptive broadcast radar system, comprising: defining a cluttercomponent for said received signal at a receiver, wherein said cluttercomponent comprises a direct path signal and a scattered signal;defining a channel transfer function; generating a sampled version ofsaid received signal according to said channel transfer function at asample time; determining a batch of data from said sampled version for asub-aperture of said receiver at said sample time; and indexing saidbatch of data into said sensor signal model.
 23. The method of claim 22,wherein said sensor signal model is a linear system model.
 24. Themethod of claim 22, wherein said batch of data includes a delay.
 25. Themethod of claim 22, further comprising linearizing a phase delay withinsaid channel transfer function to determine a doppler shift componentfor said received signal.
 26. The method of claim 25, further comprisingabsorbing said phase delay into said channel transfer function.
 27. Themethod of claim 25, wherein said phase delay correlates to a directionof clutter relative to said receiver.
 28. A method for transmitting asignal waveform from a transmitter within an adaptive broadcast radarsystem, wherein said transmitter comprises at least one sub-aperture,comprising: generating said signal waveform at said at least onesub-aperture; coding said signal waveform at said at least onesub-aperture, wherein said signal waveform is coded with transmitterdata; phase shifting said signal waveform at said at least onesub-aperture; and transmitting said coded signal waveform from an arrayelement coupled to said sub-aperture according to said phase shifting.29. The method of claim 28, further comprising applying a weight vectorto said signal waveform at said at least one sub-aperture.
 30. Themethod of claim 28, wherein said transmitter data includes the degreesof freedom associated with said transmitter.
 31. The method of claim 28,further comprising creating a train of pulses from said signal waveformwithin said transmiter, wherein said train of pulses are coded.
 32. Amethod for performing radar operations within an adaptive broadcastradar system, wherein said radar system includes a transmitter having afirst plurality of sub-apertures and a receiver having a secondplurality of sub-apertures, comprising: encoding data on a signalwaveform at SAID transmitter, wherein said data includes a number forsaid sub-apertures of said transmitter and degrees of freedom for saidtransmitter; continuously transmitting said signal waveform; determininga delay value and a doppler value for received signals at said receiver,wherein said received signals comprise direct and scattered signals ofsaid signal waveform; and regenerating a transmit signal beamcorrelating to said signal waveform from said received signals usingsaid data, said delay value and said doppler value.
 33. The method ofclaim 32, further comprising shifting a phase of said signal waveformprior to said transmitting.
 34. The method of claim 33, furthercomprising removing said phase from said received signals.
 35. Themethod of claim 32, further comprising adding a weight vector to saidsignal waveform prior to said transmitting.
 36. The method of claim 35,further comprising removing said weight vector from said receivedsignals.
 37. The method of claim 32, wherein said encoding comprisesorthogonal encoding.
 38. The method of claim 32, wherein said encodingcomprises pseudo-orthogonal encoding.
 39. The method of claim 32,further comprising generating a steering vector for said transmit signalbeam.
 40. The method of claim 32, further comprising generating a weightvector for said transmit signal beam.
 41. The method of claim 32,further comprising controlling said transmit signal beam from saidreceiver.
 42. The method of claim 32, further comprising scattering saidsignal waveform from a target to generate said scattered signals. 43.The method of claim 32, wherein said transmitter is in motion.
 44. Themethod of claim 32, wherein said receiver is in motion.
 45. The methodof claim 32, further comprising generating a data quad comprising saiddata, said delay value, and said doppler value.
 46. An adaptivebroadcast radar system, comprising: a transmitter comprising a firstplurality of sub-apertures, wherein each sub-aperture codes a signalwaveform with data; and a receiver comprising a second plurality ofsub-apertures coupled to a signal processor, wherein said signalprocessor generates a transmit beam signal according to said data withineach signal waveform.